Digital low dropout regulator

ABSTRACT

A low dropout (LDO) regulator for generating an output voltage on an output from an input voltage of an input source. The LDO regulator including a switch module to generate the output voltage. The switch module including at least two parallel connected switches responsive to corresponding switch control signals to regulate a flow of energy from the input source to the output. Each of the switches having an on-state and an off-state. A digital controller to sense the output voltage and in response to generate the switch control signals such that the output voltage is regulated to a predetermined amplitude.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.10/754,187, filed Jan. 8, 2004, which is continuation-in-part of U.S.patent application Ser. No. 10/693,787, filed Oct. 24, 2003, whichclaims the benefit of U.S. provisional application No. 60/496,957, filedAug. 21, 2003, the entire contents of which are herein incorporated byreference.

TECHNICAL FIELD

An aspect of this invention relates to power systems for electroniccircuits.

BACKGROUND

Low dropout voltage regulators are widely used to provide voltageregulation in electronics sub-systems. An LDO may be most effectivelyused when the difference voltage between the input supply voltage andthe regulated output voltage is very small. The smaller the differencevoltage is, the higher the power efficiency of the LDO. For example, theefficiency of a 2.5 volt LDO operating from a 3.3 volt supply is about75%. Although, the efficiency of an LDO is relatively poor in comparisonto a conventional high performance DC/DC converter, it may be offset bythe relatively low cost of the LDO. However, as the difference voltageacross the LDO increases, the efficiency of the LDO may becomeprohibitively low. For example, when generating a 1.2 volt regulatedsupply voltage from a 3.3 volt supply the efficiency of an LDO decreasesto a very poor 36%. Unfortunately, present day digital integratedcircuits operate with a supply voltage of approximately 1.2 volts orless, while the lowest output voltage from a typical DC/DC converter isapproximately 3.3 volts, leading to excessive losses in conventional LDOregulators.

SUMMARY

A low dropout (LDO) regulator for generating an output voltage on anoutput from an input voltage of an input source. The LDO regulatorincluding a switch module to generate the output voltage. The switchmodule including at least two parallel connected switches responsive tocorresponding switch control signals to regulate a flow of energy fromthe input source to the output. Each of the switches having an on-stateand an off-state. A digital controller to sense the output voltage andin response to generate the switch control signals such that the outputvoltage is regulated to a predetermined amplitude.

The details of one or more embodiments of the invention are set forth inthe accompanying drawings and the description below. Other features,objects, and advantages of the invention will be apparent from thedescription and drawings, and from the claims.

DESCRIPTION OF DRAWINGS

FIG. 1A is a block diagram of an aspect of a coupled inductor regulator.

FIG. 1B is a schematic diagram of an aspect of a conduction switch.

FIG. 2A is a block diagram of an aspect of a 2:1 buck regulator.

FIG. 2B is a circuit diagram of an aspect of a 2:1 buck regulator.

FIG. 2C is a representation of waveforms associated with an aspect of a2:1 buck regulator.

FIG. 3A is a block diagram of an aspect of a 1:2 boost regulator.

FIG. 3B is a circuit diagram of an aspect of a 1:2 boost regulator.

FIG. 4A is a block diagram of an aspect of a 2:−1 buck regulator.

FIG. 4B is a circuit diagram of an aspect of a 2:−11 buck regulator.

FIG. 5 is a circuit diagram of an aspect of a 1:2 boost regulator havingsynchronous rectifiers.

FIG. 6A is a schematic of an aspect of a buck configuration coupledinductor regulator having four coupled inductors.

FIG. 6B is a schematic of an aspect of a boost configuration coupledinductor regulator having four coupled inductors.

FIG. 6C is a schematic of an aspect of a flyback configuration coupledinductor regulator having four coupled inductors.

FIG. 7A is a block diagram of an aspect of a low dropout device combinedwith a 2:1 Regulator.

FIG. 7B is a schematic diagram of an aspect of a low dropout devicecombined with a 2:1 Regulator.

FIG. 8 is a block diagram of an aspect of a multi-stage regulatorsystem.

FIG. 9A is a block diagram of an aspect of buck regulator combined witha 2:1 Regulator.

FIG. 9B is a graphical representation of the current and powerdissipation in a conduction switch of a buck converter.

FIG. 10A is a block diagram of an aspect of multiple 2:1 Regulatorscombined in series with a buck regulator.

FIG. 10B is a block diagram of another aspect of multiple 2:1 Regulatorscombined in series with a buck regulator.

FIG. 11 is a block diagram of an aspect of a regulator system havingmultiple coupled inductor regulators.

FIG. 12A is a block diagram of an aspect of an amplifier system.

FIG. 12B is a block diagram of another aspect of an amplifier system.

FIG. 13 is a block diagram of an aspect of a vehicle electrical system.

FIG. 14 is a block diagram of an aspect of a power system for high speeddrivers.

FIG. 15A is a graphical representation of an aspect of a coupledinductor.

FIG. 15B is a graphical representation of an aspect of a coupledinductor.

FIG. 16A is a schematic of an aspect of a buck configuration coupledinductor regulator for generating an output voltage that isapproximately one-fourth the amplitude of the input voltage.

FIG. 16B is a timing diagram of signals corresponding to an aspect of abuck configuration coupled inductor regulator.

FIG. 16C is a schematic of an aspect of another buck configurationcoupled inductor regulator having coupled inductors.

FIG. 17A is a schematic of an aspect of an autosensing coupled inductorregulator having for automatically controlling the amplitude of theoutput voltage as a function of the input voltage.

FIG. 17B is a schematic of an aspect of an autosensing buck converterfor automatically controlling the amplitude of the output voltage as afunction of the input voltage.

FIG. 18A is a block diagram of an aspect of an LDO regulator.

FIG. 18B is a block diagram of an aspect of an LDO regulator.

FIG. 18C is a schematic diagram of an aspect of a switch array.

FIG. 18D is a schematic diagram of an aspect of a conduction switcharray.

FIG. 18E is a schematic diagram of an aspect of an conduction switcharray.

FIG. 18F is a schematic diagram of an aspect of a conduction switcharray.

FIG. 19A is a block diagram of an aspect of a power system for supplyingpower to digital logic.

FIG. 19B is a block diagram of an aspect of a conventional DC/DCconverter for generating an output voltage.

FIG. 20A is a graphical representation of voltage positioning.

FIG. 20B is a graphical representation of the output voltage and outputcurrent of a conventional DC/DC converter during an output currenttransient.

FIG. 20C is a graphical representation of the output voltage and outputcurrent of a conventional DC/DC converter with voltage positioning.

FIG. 20D is a graphical representation of the output voltage and outputcurrent of an aspect of an LDO regulator.

FIG. 21A is a block diagram of an aspect of an LDO regulator.

FIG. 21B is a schematic diagram of an aspect of a switch array.

FIG. 22 is a block diagram of an aspect of a power system including anLDO regulator for supplying power to digital logic.

FIG. 23 is a block diagram of an aspect of a power system including anLDO regulator for supplying power to digital logic.

FIG. 24 is a block diagram of an aspect of a power system including anLDO regulator for supplying power to digital logic.

FIG. 25A is a block diagram of an aspect of a multi-chip module havingan LDO regulator and digital logic.

FIG. 25B is a two-dimensional view of an aspect of a multi-chip modulehaving an LDO regulator and digital logic.

FIG. 26A is a block diagram of an aspect of a digital logic systemfabricated on a single semiconductor die.

FIG. 26B is a two-dimensional view of an aspect of a digital logicsystem fabricated on a single semiconductor die and having an LDOregulator and digital logic.

Like reference symbols in the various drawings indicate like elements.

DETAILED DESCRIPTION

FIG. 1 shows a block diagram of an aspect of a coupled-inductorregulator 10 for supplying power to one or more devices such ashigh-speed drivers and other electronic devices. The coupled inductorregulator 10 may operate open-loop to convert an input voltage, V_(DD),to a non-isolated output voltage, V_(OUT). The amplitude of the outputvoltage is approximately an integer multiple or divisor of the inputvoltage and may be determined by the configuration of thecoupled-inductor regulator 10 and the quantity of coupled inductors inthe coupled-inductor regulator 10. For example in a buck configurationhaving two coupled inductors, the coupled-inductor regulator 10 maygenerate an output voltage that is approximately one-half of the inputvoltage.

The coupled-inductor regulator 10 may include conduction switches 11,freewheeling switches 12, and two or more coupled inductors 13 arrangedin a buck, a flyback, or a boost configuration. A drive signal generator14 may generate drive signals to control the conduction switches 11. Thedrive signals are controlled to generate a total conduction timeapproaching 100%, negating a small amount of deadtime to reducecross-conduction between the conduction switches 11 and the freewheelingswitches 12.

A frequency generator 15 may generate a clock signal having an operatingfrequency. The drive signals may be synchronized to operate at theoperating frequency. In one aspect, the operating frequency may be fixedto a predetermined frequency. In another aspect, the operating frequencymay be controlled in response to changes in load conditions such asoutput current and output voltage. For example, when a change in theoutput current, such as an increase in load current, is sensed, theoperating frequency may be increased to increase the transient responseof the output. Once the coupled inductor regulator 10 has responded tothe change in load condition and has reached steady-state operatingconditions again, the operating frequency may be decreased to reducepower losses in the coupled inductor regulator 10.

The coupled inductors 13 may be tightly coupled together preferablyhaving a coefficient of coupling, K, of approximately one, where unityis the ideal value. Preferably the inductors 13 are wound together on acommon magnetic core to form an inductor assembly that provides themaximum value of coefficient of coupling. The coefficient of coupling isapproximately one being at least 0.9 and preferably greater than 0.99.The polarity for each of the windings for the coupled inductors 13 areselected so that DC currents flowing through the coupled inductors 13approximately cancel, leading to approximately zero DC current flowingthrough the magnetic core of the inductor assembly. Since there isvirtually no DC current flowing through the inductor assembly, a smallercore size may be used for the coupled inductors resulting in a smallersize (volume) and lower cost for the inductor assembly. In addition,high permeability core materials may be used for the magnetic core suchas ferrites having shapes such as bead and toroid. Lower permeabilitymaterials may also be used such as MPP cores, ferrite PQ cores, andother split core shapes.

In addition to the steady-state advantages obtained by maximizing thecoefficient of coupling between the coupled inductors, the transientresponse of the coupled inductor regulator 10 may also be improved.During a transient, the high mutual coupling between the coupledinductors may effectively cancel the inductance of the individualinductors as far as the transient load current is concerned.

FIG. 1B shows an aspect of one of the conduction switches 11 included inthe coupled inductor regulator 10. Each of the conduction switches 11may be comprised of one or more parallel switches, 16 a-16 c, that areindependently controllable. Each of the parallel switches 16 a-16 c maybe controlled by enable signals, ENB1-ENB3, to enable either all or asubset of the parallel switches 16 a-16 c. The enabled parallel switches16 a-16 c may then controlled by the same drive signal, Φ₁.

The conduction switches 11 may also be driven using a multi-level gatevoltage to reduce switching losses. For example, the amplitude of the onvoltage may be adjusted to differing predetermined levels dependent onfactors such as the current flowing through the conduction switch todecrease switching losses in the conduction switch.

FIG. 2A shows an aspect of a 2:1 regulator 20 that may be an embodimentof the coupled inductor regulator 10. The 2:1 regulator 20 may operateopen-loop to convert an input voltage, V_(DD), to a non-isolated outputvoltage that is approximately one-half the amplitude of the inputvoltage.

FIG. 2B shows a schematic diagram of an aspect of the 2:1 regulator 20.The 2:1 regulator 20 may include two buck converters operating 180degrees out of phase to generate an output voltage, Vout, from an inputvoltage. The input voltage may be a high-side voltage, V_(H), referencedto ground or to some other voltage such as a low-side voltage, V_(L).Each buck converter may include a conduction switch 22 a and 22 b, afree-wheeling switch 24 a and 24 b, and an inductor 26 a and 26 b. Anoutput capacitor 28 may filter the output voltage for each of the buckconverters. The value of the output capacitor 28 may be decreased sincethere is negligible ripple current. In addition, due to the tightcoupling between the output and the input of the 2:1 regulator 20, anycapacitance at the input works in concert with the output capacitance 28to effectively provide parallel capacitance to a load at the output.

The inductors, 26 a and 26 b, may be tightly coupled together preferablyhaving a coefficient of coupling, K, of approximately one, where unityis the ideal value. Preferably the inductors 26 a and 26 b are woundtogether on a common magnetic core to form an inductor assembly 27 thatprovides a high value of coefficient of coupling between the inductors26 a and 26 b. The polarities of the inductor windings are selected sothat the DC currents flowing through the inductors 26 a and 26 bapproximately cancel so that approximately zero DC current flows throughthe magnetic core of the inductor assembly 27. Therefore, a smaller coresize with a low permeability material may be used for the inductors 26 aand 26 b, resulting in a smaller size (volume) and lower cost for theinductor assembly 27. In addition, the transient response of the 2:1buck regulator 20 is improved due to cancellation of the individualinductances as far as transient load currents are concerned.

Any type of switches may be used for the free-wheeling switches 24 a and25 b such as synchronous rectifiers and discrete rectifiers.

Using a two level gate voltage for the conduction switches 22 a and 22 bis particularly advantageous with the 2:1 buck converter 20 since theoutput voltage, V_(DD)/2, may be used as the intermediate level voltagefor driving the conduction switches.

FIG. 2C shows waveforms associated with an aspect of the 2:1 Regulator20. Each of the conduction switches 22 a and 22 b are controlled bydrive signals that are operated approximately 180 degrees out-of-phase.The first conduction switch 22 a may be driven by a signal Φ₁, 30, thatis approximately a square-wave. The second conduction switch 22 b may bedriven a second signal Φ₁, 32, that is approximately the inverse ofsignal Φ₁, 30. A minimal amount of deadtime may be included between Φ₁and Φ ₁ to decrease any shoot-through currents that may flow from theconduction switches 22 a and 22 b through the freewheeling switches 24 aand 24 b during switching transitions. The amount of deadtime may beminimized to decrease the ripple current and to improve the transfer ofenergy to the output. When the first conduction switch 22 a isconducting, the current, I1, 34 flowing through the output inductor 26 aincreases at a linear rate. Similarly, when the second conduction switch22 b is conducting, the current, I2, 36 flowing through the outputinductor 26 b increases at a linear rate. Since the combined conductiontime of the conduction switches 22 a and 22 b approaches 100%, theamplitude of the ripple current flowing to the output capacitor 28 isnegligible, leading to a smaller output capacitor 28 for filtering theoutput.

FIG. 3A shows an aspect of a 1:2 Regulator 50 that may be an embodimentof the coupled inductor regulator 10. The 1:2 regulator 50 may operateopen-loop to convert an input voltage, V_(DD), to a non-isolated outputvoltage that is approximately twice the amplitude of the input voltage.

FIG. 3B shows a schematic diagram of an aspect of the 1:2 regulator 50.The 1:2 regulator 50 may include two boost converters operating 180degrees out of phase to generate an output voltage, Vout, from an inputvoltage. The input voltage may be a high-side voltage, V_(H), referencedto ground or to some other voltage such as a low-side voltage, V_(L).Each boost converter may include a conduction switch 52 a and 52 b, afree-wheeling switch 54 a and 54 b, and an inductor 56 a and 56 b. Anoutput capacitor 58 may filter the output voltage for each of the boostconverters. Similar to the 2:1 regulator 20, the value of the outputcapacitor 58 may be decreased since there is negligible ripple current,and due to the tight coupling between the output and the input of the1:2 regulator 50, any capacitance at the input works in concert with theoutput capacitance 58 to effectively provide parallel capacitance to aload at the output.

Each of the conduction switches 52 a and 52 b are controlled by drivesignals that are operated approximately 180 degrees out-of-phase. Thefirst conduction switch 52 a may be driven by a signal Φ₁ that isapproximately a square-wave. The second conduction switch 52 b may bedriven a second signal Φ ₁ that is approximately the inverse of signalΦ₁. A minimal amount of deadtime may be included between Φ₁ and Φ ₁ todecrease any shoot-through currents that may flow from the conductionswitches 52 a and 52 b through the freewheeling switches 54 a and 54 bduring switching transitions. The amount of deadtime may be minimized todecrease the ripple current and to improve the transfer of energy to theoutput. Since the combined conduction time of the conduction switches 52a and 52 b approaches 100%, the amplitude of the ripple current flowingto the output capacitor 58 is negligible, leading to a smaller outputcapacitor 58 for filtering the output.

The inductors, 56 a and 56 b, for each of the boost converters may betightly coupled together preferably having a coefficient of coupling, K,of approximately one. The inductors 56 a and 56 b may be wound togetheron a single magnetic core to form an inductor assembly 57 that providesa high value of coefficient of coupling. The benefits of having a highcoefficient of coupling are similar to those of the 2:1 regulator 20 andthe coupled inductor regulator 10.

Any type of switches may be used for the free-wheeling switches 54 a and55 b such as synchronous rectifiers and discrete rectifiers.

FIG. 4A shows an aspect of a 1:−1 regulator 60 that may be an embodimentof the coupled inductor regulator 10. The 1:−1 regulator 60 may operateopen-loop to convert an input voltage, V_(DD), to a non-isolated outputvoltage that is approximately the negative of the input voltage.

FIG. 4B shows a schematic of an aspect of the 1:−1 regulator 60. The1:−1 regulator 60 is similar to the 2:1 regulator 20 in function withcorresponding elements numbered in the range 60-68, except that the 1:−1regulator 60 may include two flyback regulators operating atapproximately 50% duty cycle to generate an output that is the negativeof the input voltage.

FIG. 5 shows an aspect of a coupled inductor regulator 70 that issimilar to 1:2 regulator 50 in function with corresponding elementsnumbered in the range 70-78, except that the coupled inductor regulator70 includes synchronous rectifiers 74 a and 74 b to rectify the outputsignals from the conduction switches 72 a and 72 b. The synchronousrectifiers 74 a and 74 b may advantageously reduce losses associatedwith rectifying the output signals, thereby increasing the energyefficiency of the coupled inductor regulator 70. Although thesynchronous rectifiers are illustrated as included in the boostconfiguration of the coupled inductor regulator, the synchronousrectifiers may be used as the freewheeling rectifiers in any embodimentof the coupled inductor regulator 10.

FIG. 6A shows another aspect of a coupled inductor regulator 120 forconverting an input voltage to an output voltage, Vout. The coupledinductor regulator 120 is similar to the 2:1 regulator 20 in functionwith corresponding elements numbered in the range 120-128, except thatthe coupled inductor regulator 120 includes four coupled inductors 126a-126 d having a coefficient of coupling approaching 1. Each of thecoupled inductors 126 a-126 d may be wound with a predetermined numberof turns, N1-N4 so that each of the coupled inductors may have anindividually controllable number of turns. The ratio of the turns ofeach coupled inductor to the other coupled inductors may be varied tocontrol the amplitude of the output voltage, Vout. For example, in oneaspect the turns may be set so that N1=N2=N3=N4, in which case theamplitude of the output voltage will be approximately equal toone-fourth of the input voltage. In another aspect the turns may be setso that N1=N2, N3=N4, and N1=2*N3, in which case the amplitude of theoutput voltage will be approximately equal to one third of the inputvoltage.

FIG. 6B shows another aspect of a coupled inductor regulator 130 forconverting an input voltage to an output voltage, Vout. The coupledinductor regulator 130 is similar to the 1:2 regulator 50 in functionwith corresponding elements numbered in the range 130-138, except thatthe coupled inductor regulator 130 includes four coupled inductors 136a-136 d having a coefficient of coupling approaching 1. Each of thecoupled inductors 136 a-136 d may be wound with a predetermined numberof turns, N1-N4 so that each of the coupled inductors may have anindividually controllable number of turns. The ratio of the turns ofeach coupled inductor to the other coupled inductors may be varied tocontrol the amplitude of the output voltage, Vout.

FIG. 6C shows another aspect of a coupled inductor regulator 140 forconverting an input voltage to an output voltage, Vout. The coupledinductor regulator 140 is similar to the 1:−1 regulator 60 in functionwith corresponding elements numbered in the range 140-148, except thatthe coupled inductor regulator 140 includes four coupled inductors 146a-146 d having a coefficient of coupling approaching 1. Each of thecoupled inductors 146 a-146 d may be wound with a predetermined numberof turns, N1-N4 so that each of the coupled inductors may have anindividually controllable number of turns. The ratio of the turns ofeach coupled inductor to the other coupled inductors may be varied tocontrol the amplitude of the output voltage, Vout.

FIG. 16A shows an aspect of a 4:1 regulator 150 for generating an outputvoltage, Vout, from an input voltage, Vin. The 4:1 regulator 150 mayoperate open-loop to generate Vout as a non-isolated voltage that isapproximately one-fourth the amplitude of Vin. The 4:1 regulator 150 mayinclude four drivers 152 a-152 d to buffer phase signals, φ1-φ4,corresponding to each of the drivers 152 a-152 d. The drivers 152 a-152d may be in communication with six coupled inductors 156 a-156 farranged in a lattice network.

Pairs of the coupled inductors 156 a-156 b, 156 c-156 d, and 156 e-156 fmay each be tightly coupled together preferably having a coefficient ofcoupling, K, of approximately one. Preferably each pairs of inductor 156a-156 b, 156 c-156 d, and 156 e-156 f is wound together on acorresponding common magnetic core to form inductor assemblies that mayprovide a high value of coefficient of coupling between the inductors156 a-156 b, 156 c-156 d, and 156 e-156 f. The polarities of theinductor windings are selected so that the DC currents flowing througheach pair of inductors 156 a-156 b, 156 c-156 d, and 156 e-156 fapproximately cancel so that approximately zero DC current flows throughthe magnetic core of the corresponding inductor assembly. In anotheraspect, all of the inductors 156 a-156 f may be wound on a singlemagnetic core.

The drivers 152 a-152 d may advantageously be included on a singlesemiconductor die to reduce cost, or decrease the volume of the 4:1regulator. The phase signals each may have an on-state and an off-state,and a duty cycle of approximately 25%. The phase signals may be arrangedin a timing sequence such as in one aspect, an alternating timingsequence, PS1, and in another aspect the phase signals may be arrangedin a sequential timing sequence, PS2. In the alternating timing sequencePS1, phase signals φ1-φ3-φ2-φ4 respectively are applied to the drivers152 a-152 b-152 c-152 d (see FIG. 16B). In the sequential timingsequence PS2, phase signals φ1-φ2-φ3-φ4 respectively are applied to thedrivers 152 a-152 b-152 c-152 d. The coupled inductors 156 a-156 fpreferably have a coefficient of coupling approaching 1 and may be woundwith approximately an equal number of turns on the same magnetic corestructure. An output capacitor 158 may filter the output voltage toreduce noise and ripple voltage. Similar to the 2:1 regulator 20 thevalue of the output capacitor 158 may be decreased since there isnegligible ripple current, and the capacitance at the input works inconcert with the output capacitance.

FIG. 16B shows signals and waveforms associated with an aspect of the4:1 regulator 150. The phase signals, T1-T4, show the timingrelationship between each of the phases, φ1-φ4. Each of the phases,φ1-φ4, may have a duty cycle of approximately 25% and an amplitude ofapproximately Vin. Signal PS2-A shows the waveform at node A of FIG. 16Awhen the PS2 phase sequence of the phase signals, φ1-φ4 is applied tothe drivers 152 a-152 d. Signal PS2-B shows the waveform at node B ofFIG. 16A when the PS2 phase sequence of the phase signals, φ1-φ4 isapplied to the drivers 152 a-152 d. The amplitude of signals PS2-A andPS2-B may be approximately Vin/2.

Signal PS1-A shows the waveform at node A of FIG. 16A when the PS1 phasesequence of the phase signals, φ1-φ4 is applied to the drivers 152 a-152d. Signal PS1-B shows the waveform at node B of FIG. 16A when the PS1phase sequence of the phase signals, φ1-φ4 is applied to the drivers 152a-152 d. The amplitude of signals PS1-A and PS1-B may be approximatelyVin/2. The signal frequency of signals PS1-A and PS1-B is approximatelytwice the frequency of signals PS2-A and PS2-B leading to potentiallysmaller inductance values when the PS1 timing sequence is used versusthe PS2 timing sequence.

FIG. 16C shows an aspect of a coupled inductor regulator 160 forgenerating an output voltage, Vout, from an input voltage, Vin. Thecoupled inductor regulator 160 is in a buck configuration and mayinclude multiple coupled inductors 166 arranged in a lattice networkhaving any quantity of stages ranging from 1 to N. The coupled inductorregulator 160 may operate open-loop to generate Vout as a non-isolatedvoltage that is approximately equal to Vin/(2N). The coupled inductorregulator 160 may include drivers 162 to buffer phase signals,φ₁-φ_(2N), corresponding to each of the drivers 162. Each of the phasesignals, φ₁-φ_(2N), may have a duty cycle of approximately (100/2N) %and an amplitude of Vin. The phase signals may be arranged in any timingsequence such as PS1 and PS2 as shown with the 4:1 regulator 150.

Pairs of the coupled inductors 167 within each stage may be tightlycoupled together preferably having a coefficient of coupling, K, ofapproximately one. Preferably each pair of inductors 167 is woundtogether on a corresponding common magnetic core to form inductorassemblies that may provide a high value of coefficient of couplingbetween the inductors 166. For example, stage two may have two inductorassemblies and stage three may have four inductor assemblies. Thepolarities of the inductor windings are selected so that the DC currentsflowing through each pair of inductors 167 approximately cancel so thatapproximately zero DC current flows through the magnetic core of thecorresponding inductor assembly. In another aspect, all of the inductors166 may be wound on a single magnetic core.

FIG. 7A shows an aspect of a regulator system 200 for generating aregulated output voltage, Vout, from an input voltage, Vin. Theregulator system 200 includes a low dropout regulator, LDO, 202 inseries with a coupled inductor regulator 204. The LDO 202 may controlthe LDO output, Vx, as a function of one or more feedback signals. Inone aspect, a feedback signal 206 from the output of the coupledinductor regulator 204 may communicate a sample of Vout to the LDO 202to be compared to a reference voltage. The LDO 202 may regulate the LDOvoltage, Vx, as a function of comparing Vout to the reference voltage.The coupled inductor regulator 204 may generate Vout as a function of afixed ratio of Vx. For example, if a 2:1 regulator is used for thecoupled inductor regulator 204, then Vout is approximately equal toone-half the amplitude of Vx. In another example, if a 1:2 regulator isused for the coupled inductor regulator 204, then Vout is approximatelyequal to twice the amplitude of Vx. In another aspect, multiple feedbacksignals may be used for multi-loop control of the regulator system 200.Any multi-loop control techniques may be employed such as weightedfeedback signals, selecting one feedback signal from amongst themultiple feedback signals, and varying the response time of eachfeedback loop. For example, the inner feedback loop from Vx to the LDO202 may be set slower than an outer loop from Vout to the LDO 202. Anytype of linear regulator may be used for the low drop out regulator. Inone aspect the LDO 202 and the coupled inductor regulator 204 may befabricated on a single integrated circuit 201 with separate inductorassemblies for the output inductors and coupled inductors. In anotheraspect, the order of the coupled inductor regulator 204 and the LDO 202may be reversed so that the coupled inductor regulator 204 generates theintermediate voltage, Vx, from the input voltage, Vin, and the LDO 202generates the output voltage, Vout. The LDO 202 may receive a feedbacksignal from Vout and a feedforward signal from the intermediate voltage.

FIG. 7B shows an aspect of a preferred embodiment of a regulator system210 in which a combination of a reference amplifier 214 and Field EffectTransistor 216 form an LDO having an LDO output, Vx. A 2:1 regulator 212generates an output voltage, Vout, from Vx, where Vout is approximatelyequal to one-half of the amplitude of Vx. An output capacitor 214 mayfilter the output from the 2:1 regulator 212. The regulator system 210advantageously only requires the LDO to supply a current, Ix, that isone-half the amplitude of the output current, Iout, thereby lowering thecost of the FET 204 and heat sink requirements for the LDO. A capacitor218 may also be included at the output of the FET 216 to possiblyimprove the stability of the LDO. In one exemplary regulator system 210,Vin may be 3.3V and Vout may be in the range of 1.2V to 1.5V. In anotherexemplary regulator system 210, Vin may 2.5V and Vout may be in therange of 0.8V to 1.2V.

FIG. 8 shows an aspect of a two stage regulator system 300 forgenerating an output voltage, Vout, that is approximately equal toone-fourth of the amplitude of the input voltage, Vin. A first stage 2:1regulator 302 is connected in series with a second stage 2:1 regulator304. The first stage 2:1 regulator 302 may operate at twice the voltage,but half of the current of the second stage 2:1 regulator 304. Since thetwo stages 302 and 304 operate at different voltages, the semiconductorprocesses used for each of the stages 302 and 304 may be optimized forthe operating voltage. For example, the first stage 2:1 regulator 302may be made using a 0.5 um equivalent logic transistor process, whilethe second stage 2:1 regulator 304 may be made using a 0.25 umequivalent logic transistor process. Selecting the process to be usedfor each regulator stage based on the voltage of the stage may beapplied to any of the embodiments of the coupled inductor regulator aswell as any configuration of coupled inductor regulators such as acombination of series, parallel, and tapped coupled inductor regulators.By optimizing the process that is used for each of the coupled inductorregulators, the die size may be reduced leading to a substantialdecrease in cost.

FIG. 9A shows an aspect of a very low voltage high current regulator 320(VLVHC regulator) for generating a low voltage output. The VLVHCregulator 320 may include a buck converter 322 followed by a 2:1regulator 324. In one aspect the buck converter 322 and the 2:1regulator 324 may be fabricated on a single integrated circuit 201 withseparate inductor assemblies for the output inductors and coupledinductors. The buck converter 322 may be any type of buck converter suchas a traditional buck converter having one or more output phases. One ormore feedback signals for controlling the output of the buck converter322 may be communicated to the buck converter 322 from various points inthe VLVHC regulator 320 such as the output of the 2:1 regulator 324 andthe output of the buck converter 322. Since the output current of thebuck converter 322 is half the amplitude that it would be if a singlebuck converter converted Vin directly to Vout, the output devices of thebuck converter 322 may advantageously be reduced in size by a factor ofat least 2 compared to generating Vout with only a single buckconverter. The output devices may include devices such as outputcapacitors and output inductors. The invention advantageously recognizesthat the ripple voltage at the output of the 2:1 regulator 324 will bereduced by a factor of two from the ripple voltage at the output of thebuck converter 322, therefore smaller output devices may be used for thebuck converter 322. In addition, the volumetric efficiency of capacitorsis typically directly related to the voltage rating of a capacitor sothat typically the volume of a capacitor decreases by

(V_(H)/V_(L))²

as the voltage rating of the capacitor rating increases, where VH is thehigher voltage rating and VL is the lower voltage rating. In addition,the volume of inductors is directly related to the square of the currentthat flows through an inductor, so that the volume of an inductordecreases as the maximum rated current of the inductor decreases. In oneaspect, multiple feedback signals may be used for multi-loop control ofthe VLVHC regulator 320. Any multi-loop control techniques may beemployed such as weighted feedback signals, selecting one feedbacksignal from amongst the multiple feedback signals, and varying theresponse time of each feedback loop. For example, the inner feedbackloop from the buck converter output to the buck converter 322 may be setslower than an outer loop from Vout to the buck converter 322.

In addition, the overall power efficiency of the VLVHC regulator 320 maybe lower than if only a non-isolated buck converter were used to convertVin to Vout. FIG. 9B shows a waveform 326 of current, Ids, flowingthrough a conduction switch in a non-isolated buck converter thatgenerates Vout from Vin. The duty cycle in the buck converter may beapproximately four times lower than the duty cycle in the VLVHCregulator 320. Since the duty cycle is lower, the peak current, Ids,flowing through the conduction switch may be approximately four timesgreater than the peak current flowing through the conduction switch ofthe VLVHC regulator 320, leading to switching losses in the buckconverter that may be approximately four times greater the switchinglosses in the buck converter of the VLVHC regulator 320. A waveform 328of the power dissipation, Pd, in the conduction switches of thenon-isolated buck converter shows the high switching losses occurringduring switching of the conduction switches.

FIG. 10A shows another aspect of a very low voltage high currentregulator 340 (VLVHC regulator) for generating a low voltage output. TheVLVHC regulator 340 is similar to the VLVHC regulator 320 in function,except that the VLVHC regulator 340 may include two or more coupledinductor regulators 342 and 344 following an energy source 346. Thecoupled inductor regulators 342 and 344 are preferably a buck version(2:1 regulator, 3:1 regulator, 4:1 regulator, etc.) of the coupledinductor regulator to take advantage of the decrease in size of theoutput devices of the energy source 346, however the scope of theinvention includes any embodiment of the coupled inductor regulator suchas a boost regulator having a 1:2 boost ratio.

FIG. 10B shows another aspect of a very low voltage high currentregulator 370 (VLVHC regulator) for generating a low voltage output. TheVLVHC regulator 370 is similar to the VLVHC regulator 340 in function,except that the VLVHC regulator 370 may include two or more coupledinductor regulators 372 and 374 preceding an energy source 376. Thecoupled inductor regulators 372 and 374 may be any configuration such asa buck configuration (2:1 regulator, 3:1 regulator, 4:1 regulator,etc.), a boost configuration (1:2 regulator), and a flybackconfiguration (1:−1 regulator) of the coupled inductor regulator.

FIG. 11 shows an aspect of a multi-stage regulator system 350 forconverting an input voltage, Vin, to several output voltages,Vout1-Vout3. The multi-stage regulator system 350 may include an energysource 352 such as a buck converter connected to two or more coupledinductor regulators 354-360. The multi-stage regulator system 350 mayadvantageously generate several intermediate voltages such asVout1-Vout3, while minimizing the cost of the overall implementation byreducing the cost of the output devices of the energy source 352.

FIGS. 12A and 12B show aspects of single supply amplifier systems 400and 410 for powering a load such as a speaker 402 from a power amplifier404. Conventional single supply amplifier systems include a large DCblocking capacitor in series with the load and power amplifier to removeany DC components from the signal from the signal that drives the load.The single supply amplifier system 400 advantageously may use a coupledinductor regulator to generate a second supply voltage, V_(o2), from thefirst supply voltage V_(DD). The second supply voltage may be used toeliminate DC voltage from appearing across the load 402.

In one aspect, a 2:1 regulator 406 may generate a voltage, V_(o2), thatis approximately one-half of the amplitude of V_(DD). The voltage,V_(o2), may be applied to one end of the load 402 to bias the load 402so that no DC voltage appears across the load 402, thereby eliminatingthe need for a DC blocking capacitor. In one aspect the 2:1 regulator406 and the power amplifier 404 may be fabricated on a single integratedcircuit 401.

In another aspect, a 1:−2 regulator 408 may generate a voltage, −V_(DD),that is the negative of the high side supply voltage, V_(DD), for thepower amplifier 404. The voltage −V_(DD) is used as the low side supplyvoltage for the power amplifier 404 to convert the power amplifier 404into a dual power supply amplifier. The power amplifier 404 may thengenerate generates an output that is approximately centered about zerovolts and has approximately no DC component, thereby eliminating theneed for a DC blocking capacitor. In one aspect the 1:−1 regulator 408and the power amplifier 404 may be fabricated on a single integratedcircuit 411.

FIG. 13 shows a vehicle electrical system 500 that may be powered by a24 volt battery 502. The vehicle electrical system 500 may include amixture of both 24 volt loads 504 and 12 volt loads 506. For example,the basic electrical systems of an automobile such as the engine,compressor, fan, lights, and air conditioning may all be driven from the24 volt battery 502. Whereas, one or more accessories for the automobilesuch as a stereo, computer, cigarette charger, and global positioningsystem may have been designed for a 12 volt automotive system andtherefore require a 12 volt supply to provide power. A 2:1 regulator 508may provide a low cost source of 12 volt power derived from the 24 voltbattery 502 so that legacy automotive accessories that require 12 voltpower may be employed in the automotive electrical system 500.

FIG. 17A shows an aspect of an autosensing regulator 510 for generatingan output voltage, Vout, from an input voltage, Vin. The autosensingregulator 510 is particularly suitable for being used as the 2:1regulator 508 in the automotive electrical system 500. The autosensingregulator 510 may include an autosensor 530 to sense the input voltageand control a 2:1 regulator 520. The 2:1 regulator may be similar to the2:1 regulator 20 in function with corresponding elements numbered in therange 520-528. The autosensor 530 may automatically control the 2:1regulator 520 as a function of the amplitude of the input voltage. Forexample, when Vin is greater than a predetermined voltage level, theautosensor 530 may set the duty cycle of the conduction switches 522 aand 522 b to 50% each so that the 2:1 regulator 520 may generate anoutput voltage that is approximately one-half the amplitude of Vin, andwhen Vin is less than the predetermined voltage level, the autosensor530 may set the conduction switches 522 a and 522 b both to thecontinuous on-state so that Vout is approximately equal to Vin.

FIG. 17B shows another aspect of an autosensing regulator 540 forgenerating an output voltage, Vout, from an input voltage, Vin. Theautosensing regulator 540 may include an autosensor 560 to sense theinput voltage and control a conventional buck converter 550 as afunction of the amplitude of the input voltage. Any type of buckconverter 552 may be employed. The buck converter 550 may include aconduction switch 552, a free-wheeling diode 554, an output inductor556, and an output capacitor 558. In one aspect, the autosensor may setthe conduction switch 552 to the on-state continuously if Vin is lessthan a predetermined voltage level, and if Vin is greater than thepredetermined voltage level the autosensor 560 may enable the conductionswitch to be driven by a variable duty cycle signal to maintain aconstant output voltage.

FIG. 14 shows a driver power system 600 for supplying power to a highspeed line driver assembly (DDR) 602. A 2:1 regulator 604 may generateV_(TT) voltage from the V_(DDQ) voltage. The V_(TT) voltage isapproximately one-half of the V_(DDQ) voltage. The 2:1 regulator 604advantageously may use output filter capacitors that are much smallerthan conventional regulators may require. The V_(TT) voltage may supplypower to termination devices 606 a and 606 b and the DDR 602. Tocompensate for a V_(TT) voltage that is not precisely one-half of theV_(DDQ) voltage, a reference voltage, V_(REF), 608 for the DDR 602 maybe derived from the VTT voltage. In addition, a filter 610 may filterthe reference voltage 608 to attenuate noise components.

FIG. 15A shows an aspect of a coupled inductor 700 wound on a torroid.The windings of the coupled inductor 700 are arranged so that DCcurrents flowing through the windings cancel. By minimizing the combinedDC current flowing in the coupled inductor 700, saturation of thetorroid is prevented and high permeability materials such as ferritesmay used for the torroid to reduce core losses.

FIG. 15B shows another aspect of a coupled inductor 710 wound on aplanar assembly. The coupled inductor 710 is similar in function to thecoupled inductor 700 such as the windings are arranged so that DCcurrents flowing through the windings cancel, so that high permeabilitymaterials such as ferrites may be used for the core.

FIG. 18A shows an aspect of a digital low dropout (LDO) regulator 900for generating a regulated output voltage, Vout, from an input voltage,Vin. The digital LDO regulator 900 may include a switch module 902 thatis controlled by a digital controller 904. The switch module 902 may beimplemented as either a coupled inductor regulator or as a digitalresistor. The coupled inductor regulator may be any configurationdescribed in this specification such as buck, boost, and flybackconfigurations. The digital resistor comprises a switch array withoutany coupled inductors. In each case, a switch array 906 included in theswitch module 902 is controlled by the digital controller to control theregulated output voltage. FIG. 18C shows an aspect of a switch array950. Control signals, CNTL1-CNTLX, may independently control theswitches 952 on a cycle-by-cycle basis. The switches 952 may becontrolled in groups and individually. In one aspect, the switch array906 may be implemented as the switch array 1006 shown in FIG. 18D.

FIG. 18B shows an aspect of a coupled inductor type of digital LDOregulator 1000 for generating a regulated output voltage, Vout, from aninput voltage, Vin. The digital LDO regulator 1000 may include a coupledinductor regulator 1002 that is controlled by a digital controller 1004.The coupled inductor regulator 1002 may be any configuration such asbuck, boost, and flyback. At least one of the conduction and flybackswitches of the coupled inductor regulator 1002 is implemented as aswitch array 1006 such as a conduction switch array, a flyback switcharray, and combinations thereof. The switch array 1006 may include anarray of switches that are controllable on a cycle-by-cycle basis ingroups of one or more switches. Any type of switches may be used for theconduction switch array 1006 such as MOSFETs, NMOS, PMOS, and BJTsalthough an array of MOSFETs on a single integrated circuit ispreferable. By controlling the quantity of switches that conduct energyduring a conduction cycle the power losses in the conduction switcharray may be controlled. For example, if the conduction switch array1006 is implemented as an array of MOSFETs, the voltage drop across theconduction switch array 1006 may be controlled as a function of thequantity of MOSFETs that conduct energy during a cycle.

FIG. 18D shows an aspect of an exemplary coupled inductor regulator 1002having a 2:1 buck configuration. The buck configuration coupled inductorregulator 1020 may include one or more conduction switch arrays 1022 aand 1022 b to replace the conduction switches that are described in anearlier portion of this specification. In the buck configuration, theconduction switch arrays 1022 a and 1022 b may receive a high-sidevoltage such as V_(DD). Preferably each conduction switch of the coupledinductor regulator 1002 is replaced by a conduction switch array 1022,however it is within the scope to only replace a subset of theconduction switches with conduction switch arrays 1022. For example, ina 1:4 boost configuration of the coupled inductor regulator 1002, oneconduction switch may be replaced with a conduction switch array, whilethe remaining three conduction switches are not replaced by conductionswitch arrays. Control signals, CNTLxx, may independently control theswitches 1030 a and 1030 b that comprise each of the conduction switcharrays 1022 a and 1022 b. Freewheeling switches 1024 a and 1024 b mayconnect to each of the conduction switch arrays 1022 a and 1022 b. Eachof the freewheeling switches 1024 a and 1024 b in the buck configurationmay receive a low-side voltage such as V_(L). Coupled inductors 1028 aand 1028 b may connect between the conduction switch arrays 1022 a and1022 b and the output. An output capacitor 1029 may connect to theoutput.

FIG. 18E shows another aspect of the exemplary coupled inductorregulator 1002. Here, a buck configuration coupled inductor regulator1040 may include one or more freewheeling switch arrays 1044 a and 1044b to replace the freewheeling switches that are described in an earlierportion of this specification. In the buck configuration, thefreewheeling switch arrays 1044 a and 1044 b may receive a low-sidevoltage such as V_(L). Preferably each freewheeling switch of thecoupled inductor regulator 1002 is replaced by a freewheeling switcharray 1044, however it is within the scope to only replace a subset ofthe freewheeling switches with freewheeling switch arrays 1044. Forexample, in a 1:4 boost configuration of the coupled inductor regulator1002, one freewheeling switch may be replaced with a freewheeling switcharray, while the remaining freewheeling switches are not replaced byfreewheeling switch arrays. Control signals, CNTLxx, may independentlycontrol the switches 1050 a and 1050 b that comprise each of thefreewheeling switch arrays 1044 a and 104 b. Conduction switches 1042 aand 1042 b may connect to each of the freewheeling switch arrays 1044 aand 1044 b. Each of the conduction switches 1042 a and 1042 b in thebuck configuration may receive a high-side voltage such as V_(DD).Coupled inductors 1048 a and 1048 b may connect between the freewheelingswitch arrays 1044 a and 1044 b and the output. An output capacitor 1049may connect to the output.

FIG. 18F shows another aspect of the exemplary coupled inductorregulator 1002. Here, a buck configuration coupled inductor regulator1060 may include one or more conduction switch arrays 1062 a and 1062 b,and freewheeling switch arrays 1064 a and 1064 b to replace theconduction switches and freewheeling switches that are described in anearlier portion of this specification. In the buck configuration, theconduction switch arrays 1062 a and 1062 b may receive a high-sidevoltage such as V_(DD), and the freewheeling switch arrays 1064 a and1064 b may receive a low-side voltage such as V_(L). Preferably eachconduction switch and each freewheeling switch of the coupled inductorregulator 1002 are replaced respectively by a conduction switch array1062 and a freewheeling switch array 1064, however it is within thescope to only replace a subset of the conduction switches andfreewheeling switches with conduction switch arrays 1062 or freewheelingswitch arrays 1064. For example, in a 1:4 boost configuration of thecoupled inductor regulator 1002, one conduction switch may be replacedwith a conduction switch array, while the remaining conduction switchesand freewheeling switches are not replaced by switch arrays. Controlsignals, CNTLxx, may independently control the switches 1070 a, 1070 b,1071 a, and 1071 b that comprise each of the conduction switch arrays1062 a and 1062 b, and freewheeling switch arrays 1064 a and 1064 b.Coupled inductors 1068 a and 1068 b may connect between the switcharrays 1062 a, 1062 b, 1064 a and 1064 b and the output, Vout. An outputcapacitor 1069 may connect to the output.

Any combination of switch arrays may be used in the coupled inductorregulator 1002 so long as at least one conduction switch or freewheelingswitch is replaced by a switch array. For example, the buckconfiguration coupled inductor regulators 1020, 1040, and 1060 shown inFIGS. 18D, 18E, and 18F merely show 3 of the 15 possible combinations ofswitch arrays for a 2:1 buck configuration. Further combinations arepossible for the flyback and boost configurations as well as increasedorder configurations.

Referring to FIG. 18B, the digital controller 1004 may, as a function ofthe output voltage, generate the control signals to control theconduction switch array 1006. An analog-to-digital converter (ADC) 1010may generate a digital signal that corresponds to the output voltage.Any type of ADC may be employed. A digital circuit 1012 such as adigital signal processor may process the digital signal to determine thestate of the control signals for regulating the output voltage to apredetermined level. The control signals may control the quantity ofconducting switches in the conduction switch array 1006 to control thevoltage drop across the conduction switch array 1006 and thereby theoutput voltage.

In one exemplary digital LDO regulator having an input voltage of 3.3volts, the coupled inductor regulator 1002 may be a buck configurationso that the output voltage would be approximately equal to 1.65 volts(one-half of the input voltage) if the coupled inductor regulator 1002were operated open-loop. However, with the digital controller 1004controlling the conduction switch array 1006, the output voltage may beregulated at voltages approximately equal to or less than 1.65 volts.

In another aspect, the digital circuit 1012 may include a burst mode1012 for controlling the duty cycle of the switches of the conductionswitch array 1006. Burst mode 1012 may advantageously provide improvedregulation during operation at low output current, Iout, levels such aswhen Iout is so low that controlling the quantity of conducting switchesis not sufficient to regulate the output voltage. The burst mode may beoperated in combination with controlling the quantity of conductingswitches in the conduction switch array 1006.

FIG. 19A shows an aspect of a power system 1100 for supplying aregulated supply voltage to digital logic 1102 such as a centralprocessing unit (CPU). The power system 1100 may include a DC/DCconverter 1104 followed by a digital LDO regulator 1106 to generate theregulated supply voltage. Any type of DC/DC converter 1104 may be usedsuch as a single phase output and a multi-phase output. The fastresponse time of the digital LDO regulator 1106 advantageously may makethe digital LDO regulator 1106 suitable for supplying power directly tothe digital logic 1102.

The digital LDO regulator 1106 may include a coupled inductor regulator1108 and digital controller 1110 as described earlier in thisspecification. The digital controller 1106 may include a control loop,1112 a and 1112 b, to control the output voltage of the DC/DC converter1104 so that the input voltage of the coupled inductor regulator 1108 isminimized, thereby limiting the power dissipation and increasing themaximum output current that the digital LDO regulator 1108 may supplywithout dropping out of regulation during a transient load change. Thecontrol loops of the digital controller 1110 may be configured so thatthe control loop of the coupled inductor regulator 1108 is a fast loop1112 a while the control loop of the DC/DC converter 1104 is a slow loop1112 b. Since the DC/DC converter 1104 does not require a fast responsetime, the DC/DC converter 1104 may advantageously be operated at a loweroperating frequency leading to higher power efficiency. The conductionlosses are decreased and the efficiency of the DC/DC converter 1004 isfurther increased since the output current of the DC/DC converter 1004is less than one-half of the output current of a DC/DC converter used ina conventional power system. Additionally, the power system does notrequire voltage positioning of the output voltage. Conventional DC/DCconverters that supply power to a CPU typically include voltagepositioning to reduce the voltage stress applied to the CPU duringtransient load currents. Voltage positioning is a technique wherein theamplitude of the output voltage is programmed to vary as a function ofthe amplitude of the output current.

FIG. 19B shows a block diagram of an aspect of the DC/DC converter 1104.A power stage 1120 may generate a chopped output from an input voltage,Vin. An output filter 1122 may filter the chopped output to generate aDC output, Vout. A control circuit 1124 may control the power stage as afunction of the DC output.

FIG. 20A shows voltage positioning of an output voltage. A firstwaveform 1150 shows an output voltage of a conventional converter thatis programmed to have a negative slope with increasing output current. Asecond waveform 1152 shows an aspect of a digital LDO regulator whereinthe output voltage has a flat slope for all values of output current. Athird waveform 1154 shows another aspect of a digital LDO regulatorwherein the output voltage has a positive slope with increasing outputcurrent.

FIG. 20B shows waveforms of the output voltage, Vout, 1160 and outputcurrent, Iout, 1162 of a conventional DC/DC converter that does notinclude voltage positioning. The output voltage may fluctuate duringtransients in the output current due to the relatively slow response ofthe conventional DC/DC converter. The fluctuation of Vout during thehigh-to-low transition of the output current may be especiallyproblematic due to possible voltage overstress of the CPU.

FIG. 20C shows waveforms of the output voltage, Vout, 1170 and outputcurrent, Iout, 1172 of a conventional DC/DC converter that includesvoltage positioning. The amplitude of the output voltage at high outputcurrents may be programmed to be lower than the amplitude of the outputvoltage at low output currents. At high output current amplitudes, theamplitude of the output voltage is programmed to be less than the steadystate value without voltage programming. During a high-to-low outputcurrent transient, the maximum overshoot of the output voltage isreduced since the overshoot begins from a lower steady-state outputvoltage amplitude. The lower maximum overshoot reduces the voltagestress on the digital circuit that receives power. However, voltagepositioning may have a disadvantage at lower output current levels,wherein the higher output voltage level may lead to an increase inleakage current in the digital circuit. As CMOS device geometry isscaled below 0.13 u a slight increase in supply voltage may cause anexponential increase in leakage current in the digital circuit. Yet,voltage positioning results in a higher supply voltage when the digitalchip operates at lower speeds (low supply current), leading to highleakage when the digital circuit is supposed to be relatively inactive.In addition, since voltage positioning causes the lowest supply voltagelevel to be applied to the digital circuit when the supply current ishighest, the maximum clock frequency of the digital circuit may belimited.

FIG. 20D shows waveforms of the output voltage, Vout, 1180 and outputcurrent, Iout, 1182 of an aspect of a digital LDO regulator. The fastresponse time of the LDO regulator negates the need for voltagepositioning, since voltage transients during load current changes arenegligible. Since the output voltage to the digital circuit during highsupply current operation may be controlled to be higher than undervoltage positioning, the digital circuit may operate at a higher maximumclock frequency.

FIG. 21A shows an aspect of a digital LDO regulator 1200 configured as adigital resistor for generating a regulated output voltage, Vout, froman input voltage, Vin. The digital LDO regulator 1200 is similar infunction to digital LDO regulator 1000 with corresponding elementsnumbered in the range 1200-1208, except that digital LDO regulator 1200includes a switch array 1202 instead of a coupled inductor regulator forregulating the output voltage. The digital resistor configured LDOregulator 1200 may regulate the output voltage to any level that is lessthan the input voltage. A digital controller 1204 operates in a similarmanner to the digital controller 1004 of the digital LDO regulator 1000to control the switch array 1202. An output capacitor 1220 may filterthe output voltage to provide energy storage and noise suppression.

FIG. 21B shows an aspect of the switch array 1202 for the digital LDOregulator 1200. Any type of switches may be used for the switch array1202 such as MOSFETs, NMOS, PMOS, and BJTs although an array of MOSFETson a single integrated circuit is preferable. By controlling thequantity of switches that conduct energy during a conduction cycle thepower losses in the switch array 1202 may be controlled. For example, ifthe switch array 1202 is implemented as an array of MOSFETs, the voltagedrop across the switch array 1202 may be controlled as a function of thequantity of MOSFETs that conduct energy during a cycle. Control signals,CNTL1-CNTLX, may independently control the switches 1208 on acycle-by-cycle basis. The switches 1208 may be controlled in groups andindividually.

FIG. 22 shows an aspect of a power system 1300 for supplying power to aCPU 1302. The power system 1300 may include a digital LDO regulator 1304configured as a digital resistor type. The digital LDO regulator 1304may generate a regulated output voltage, Vout, from an input voltage,Vin. A DC/DC converter 1306 may generate the input voltage for thedigital LDO regulator 1304. The input voltage is preferably The digitalLDO regulator 1304 is similar in function to digital LDO regulator 1200.

FIG. 23 shows an aspect of a power system 1400 for supplying power to aCPU 1402. The power system 1400 may include a digital LDO regulator 1404configured as a coupled inductor type for regulating the power to theCPU 1402. The digital LDO regulator 1404 may include two or more coupledinductor regulators 1406 a and 1406 b in series to generate an outputvoltage, Vout, from an input voltage, Vin. A DC/DC converter 1408 maygenerate the input voltage for the digital LDO regulator 1404. Thedigital LDO regulator 1404 may include a digital controller 1410 tocontrol the coupled inductor regulators 1406 a and 1406 b, and the DC/DCconverter 1408. The digital controller 1410 may function in a similarmanner to digital controller 1004 of LDO regulator 1000 with theadditional function of controlling the output voltage of the DC/DCconverter 1408. The digital controller 1410 may control the outputvoltage of the DC/DC converter 1408 either directly via a feedbacksignal from the output of the DC/DC converter 1408 or indirectly viaother circuit characteristics such the current flowing to the CPU andthe output voltages of the coupled inductor regulators.

FIG. 24 shows an aspect of a power system 1500 for supplying power to aCPU 1502. The power system 1500 may include a digital LDO regulator 1504for generating a regulated output voltage, Vout, from an input voltage,Vin. A DC/DC converter 1508 may generate the input voltage for thedigital LDO regulator 1504. The digital LDO regulator 1504 may include aswitch module 1506 that is configured for operation as either a coupledinductor type or a digital resistor type. For coupled inductor operationthe switch module is a coupled inductor regulator similar in function tothe coupled inductor regulator 1006. For digital resistor operation theswitch module 1506 is a switch array similar in function to switch array1202. The digital LDO regulator 1504 may include an ADC 1510 and a DSP1512 to control the switch module 1506 as a function of the outputvoltage and external stimuli such as CPU control signals and othercircuit signals. The CPU control signals may include wake mode, burstmode, and sleep mode as well as any other CPU control signals.

FIG. 25A shows an aspect of a multi-chip module 1600. The multi-chipmodule 1600 may include digital logic 1602 to provide any type offunction. The digital logic 1602 may be any type of logic such ascomplex logic and processors such as a Power PC and X86 chips. A digitalLDO regulator 1604 in accordance with the teachings of thisspecification may receive an input voltage and generate a regulatedoutput voltage therefrom to power the digital logic 1602.

FIG. 25B shows a two-dimensional side view of an aspect of themulti-chip module 1600. Any type of physical configuration may be usedfor the multi-chip module 1600. A heatsink 1606 may provide a thermalpath for heat generated by the digital logic 1602 and the LDO regulator1604. A package substrate 1612 may interconnect the digital logic 1602to the digital LDO regulator 1604. One or more pins 1608 may connect tothe package substrate 1612 to provide an interconnect from themulti-chip module 1600 to another assembly such as a motherboard. For acoupled inductor configuration of the LDO regulator 1604, one or morecoupled inductors 1610 associated with the digital LDO regulator 1604may be mounted on the package substrate 1612.

FIG. 26A shows an aspect of a semiconductor device 1700 fabricated on asingle semiconductor die 1701. The semiconductor device including adigital LDO regulator 1702 and a digital circuit 1704. The semiconductordevice 1700 may include digital logic 1702 to provide any type offunction. The digital logic 1702 may be any type of logic such ascomplex logic and processors such as a Power PC and X86 chips. A digitalLDO regulator 1704 in accordance with the teachings of thisspecification may receive an input voltage and generate a regulatedoutput voltage therefrom to power the digital logic 1702

FIG. 26B shows a two-dimensional side view of an aspect of thesemiconductor device 1700. A heatsink 1706 may provide a thermal path todissipate heat generated on the single semiconductor die 1701. Aninterconnect assembly 1712 may interconnect the semiconductor die 1700to other devices. Any type of interconnect assembly 1712 may be usedsuch as a motherboard. One or more pins 1708 may connect to theinterconnect assembly 1712. For a coupled inductor configuration of theLDO regulator 1704, one or more coupled inductors 1710 associated withthe LDO regulator 1704 may be mounted on the interconnect assembly 1712.

A number of embodiments of the invention have been described.Nevertheless, it will be understood that various modifications may bemade without departing from the spirit and scope of the invention.Accordingly, other embodiments are within the scope of the followingclaims.

1. A digital low dropout (LDO) regulator for generating an output voltage on an output from a source of an input voltage, comprising: a switch module to generate the output voltage, the switch module including at least two parallel connected switches responsive to corresponding switch control signals to regulate a flow of energy from the input voltage source to the output, each of the switches having an on-state and an off-state; and a digital controller to sense the output voltage and in response to generate the switch control signals such that the output voltage is regulated to a predetermined amplitude. 